4-phase buck-boost converter

ABSTRACT

A system has an input and an output. The system includes a voltage converter, including first transistor coupled to the input and to a first switching node, second transistor coupled to the first switching node and to ground, third transistor coupled to a second switching node and to the output, fourth transistor coupled to the second switching node and to ground, and an inductor having a first terminal coupled to the first switching node and a second terminal coupled to the second switching node. The system also includes a controller coupled to the voltage converter, the controller including a state machine and a plurality of drivers to control the transistors of the voltage converter. The state machine is adaptable to cause the second and fourth transistors to conduct and the first and third transistors not to conduct, in response to current through the inductor being less than a current threshold.

CROSS-REFERENCE TO RELATED APPLICATIONS

The present application claims priority to U.S. Provisional PatentApplication No. 62/903,460, which was filed Sep. 20, 2019, is titled“4-Phase Non-Inverting Buck-Boost Converter,” and is hereby incorporatedherein by reference in its entirety.

SUMMARY

In accordance with at least one example of the disclosure, a system hasan input terminal and an output terminal. The system includes a voltageconverter, including a first transistor coupled to the input terminaland to a first switching node, a second transistor coupled to the firstswitching node and to ground, a third transistor coupled to a secondswitching node and to the output terminal, a fourth transistor coupledto the second switching node and to ground, and an inductor having afirst terminal coupled to the first switching node and a second terminalcoupled to the second switching node. The system also includes acontroller coupled to the voltage converter, the controller including astate machine and a plurality of drivers to control the transistors ofthe voltage converter. The state machine is adaptable to cause thesecond and fourth transistors to conduct and the first and thirdtransistors not to conduct, in response to current through the inductorbeing less than a current threshold.

In accordance with another example of the disclosure, a controller isfor a voltage converter having a first switching node, a secondswitching node and an inductor coupled between the first and secondswitching nodes. The controller includes a state machine and a pluralityof drivers, each driver coupled to a transistor. The state machine, inresponse to a current through the inductor being less than a firstcurrent threshold, is adaptable to transition to a state in which: afirst transistor between the first switching node and an input terminalis not conducting; a second transistor between the first switching nodeand a ground node is conducting; a third transistor between the secondswitching node and an output terminal is not conducting; and a fourthtransistor between the second switching node and the ground node isconducting.

In accordance with yet another example of the disclosure, a method forcontrolling a voltage converter includes detecting, by a controller, acurrent through an inductor coupled to a first switching node and to asecond switching node of the voltage converter. The method alsoincludes, in response to the current through the inductor being lessthan a current threshold, the controller causing a first transistorbetween the first switching node and an input terminal not to conduct,the controller causing a second transistor between the first switchingnode and a ground node to conduct, the controller causing a thirdtransistor between the second switching node and an output terminal notto conduct, and the controller causing a fourth transistor between thesecond switching node and the ground node to conduct.

BRIEF DESCRIPTION OF THE DRAWINGS

For a detailed description of various examples, reference will now bemade to the accompanying drawings in which:

FIG. 1 shows a schematic diagram of a voltage converter and a controllerin accordance with various examples;

FIG. 2 shows a schematic diagram of the voltage converter in a boost-onphase in accordance with various examples;

FIG. 3 shows a schematic diagram of the voltage converter in aboost-off, buck-on phase in accordance with various examples;

FIG. 4 shows a schematic diagram of the voltage converter in a buck-offphase in accordance with various examples;

FIG. 5 shows a schematic diagram of the voltage converter in a pausephase in accordance with various examples;

FIG. 6 shows a state diagram of operation of the voltage converter inaccordance with various examples;

FIG. 7 shows a waveform of inductor current as a function of time forthe voltage converter operating in a buck mode in accordance withvarious examples;

FIG. 8 shows a waveform of inductor current as a function of time forthe voltage converter operating in a boost mode in accordance withvarious examples;

FIG. 9 shows a waveform of inductor current as a function of time forthe voltage converter operating in a buck-boost mode in accordance withvarious examples;

FIG. 10 shows a block diagram of a system in accordance with variousexamples;

FIG. 11 shows a state diagram of operation of a controller for a voltageconverter in accordance with various examples;

FIG. 12 shows a waveform of output voltage and inductor current as afunction of time for the voltage converter operating in the buck mode inaccordance with various examples;

FIG. 13 shows a waveform of output voltage and inductor current as afunction of time for the voltage converter operating in the buck-boostmode in accordance with various examples; and

FIG. 14 shows a waveform of output voltage and inductor current as afunction of time for the voltage converter operating in the boost modein accordance with various examples.

DETAILED DESCRIPTION

Direct current (DC)-DC converters may be implemented as switched modepower supplies (SMPS). DC converters may be used in a variety ofcircuits to provide a DC output signal by converting a DC input signal.For example, DC converters may be used in systems in which power issupplied by a battery to a load, particularly where the battery voltagemay change over time (e.g., as the battery is depleted). Examples ofsuch systems include automotive applications, personal electronicdevices, Internet of Things (IoT)-connected devices, or otherbattery-powered applications. The input and output signals can havesimilar or opposite polarities. SMPS converters include buck, boost,buck-boost, and other types. Buck DC-DC converters are operable toprovide output voltages (VOUT) equal to or below the voltage of theinput signal (VIN). Boost converters are operable to provide VOUT thatis greater than or equal to VIN. Buck-boost converters provide thefunctionality of a buck converter and a boost converter. Buck-boostconverters include one or more inductors. The series inductor isenergized by the input signal and is subsequently de-energized toprovide the output signal.

An issue with buck-boost converters occurs when VOUT is approximatelyequal to VIN, referred to as the buck-boost transfer region. In thebuck-boost transfer region, buck-boost converters suffer from regulationissues related to toggling between buck mode and boost mode, whichresults in glitches such as sub-harmonic oscillations on VOUT. Inparticular, buck-boost converters have different transfer functions foroperation in buck mode (VOUT<VIN) and boost mode (VOUT>VIN). The buckmode transfer function is given by VOUT=VIN*D(buck), where D(buck) is aduty cycle value for buck mode operation in the range of 0 to 1. Theboost mode transfer function is given by VOUT=VIN/(1−D(boost)), whereD(boost) is a duty cycle value for boost mode operation in the range of0 to 1. Thus, in the buck-boost mode of operation, where VOUT=VIN, thebuck mode duty cycle, D(buck), is close to 1 and the boost mode dutycycle, D(boost), is close to 0. However, due to minimum on- andoff-times, D(buck) can approach the value of 1 (but may not be able toreach it) and D(boost) can approach the value of 0 (but may not be ableto reach it). This condition leads to the regulation issues, includingsub-harmonic oscillations on VOUT, because the transfer function is notdefined in this region.

Regulation bandwidth of a voltage converter refers to the response timeof a control loop for the voltage converter to a change in an inputcondition (e.g., load current) for the control loop. A greaterregulation bandwidth results in a shorter response time to a change inthe input condition and a lesser regulation bandwidth results in alonger response time to a change in the input condition. In boost mode,the regulation bandwidth of buck-boost converters is limited by theright-half-plane (RHP) zero frequency. The RHP zero frequency limitsregulation bandwidth because the RHP zero acts as a pole that providesgain boosting in the feedback path. As a result, the regulationbandwidth should be, for example, three to five times lower than the RHPzero frequency to avoid oscillation. The limitation on regulationbandwidth by the RHP zero frequency results in a diminished transientresponse in boost mode because the control loop reacts more slowly tochanges in the input condition (e.g., load current) to regulate VOUT ofthe voltage converter.

Example embodiments (which include a controller) address the foregoingissues for a voltage converter, such as a buck-boost, DC-DC voltageconverter. In an example, the controller includes a state machineconfigured to control transistors of the voltage converter to operate ina pause phase, in which an inductor of the voltage converter is shorted,allowing energy to be preserved in the voltage converter during thepause phase. Thus, in addition to the inductor being energized by VINand de-energized to provide VOUT, examples of this disclosure includecontrolling the voltage converter to include a phase in which energy ispreserved in the voltage converter as will be discussed below.

As a result, the transfer of input signal energy to output signal energyby the voltage converter can be regulated by altering the length of thepause phase, in which energy is preserved in the voltage converter. Thisreduces the above-mentioned problems with regulation in the buck-boostmode of operation, such as sub-harmonic oscillations, by providing aphase of operation for the voltage converter in which the inductor isneither being energized nor de-energized. Additionally, the pause phaseallows energy transfer from the voltage converter to a load to bestopped without consequently energizing the inductor, which avoids VOUTovershoots, and thus addresses the above-mentioned RHP zero limitationson bandwidth. These benefits are described more fully below withreference to various examples and the accompanying figures.

FIG. 1 depicts a system 100 in accordance with various examples. Thesystem 100 includes a voltage converter 101 and a controller 103 coupledto the voltage converter 101. In this example, the voltage converter 101is a buck-boost converter that converts an input voltage (VIN) at aninput terminal 105 to an output voltage (VOUT) at an output terminal109. The voltage converter 101 is configured to operate in multiplemodes (e.g., buck mode, boost mode, or buck-boost mode). The voltageconverter 101 includes at least a first switch 102, a second switch 104,a third switch 106, a fourth switch 108, and an inductor 110. In oneexample, the switches 102, 104, 106, 108 are transistors, such asfield-effect transistors (such as n-type or p-type metal-oxide-siliconfield-effect transistors, MOSFETs) or bipolar transistors, and arereferred to as transistors below. In the example of FIG. 1, the voltageconverter 101 also includes an input capacitor 112 and an outputcapacitor 114.

In particular, the input capacitor 112 is coupled between input terminal105 and a ground terminal 107. The first transistor 102 is also coupledto the input terminal 105 and a first switching node SW1, while thesecond transistor 104 is coupled to the first switching node SW1 and tothe ground terminal 107. For example, the drain of transistor 102 (if itis a pMOS device) is coupled to input terminal 105 and the source oftransistor 102 is coupled to the switching node SW1. Similarly, thedrain of transistor 104 (if it is a pMOS device) is coupled to theswitching node and the source of transistor 104 is coupled to ground107. The inductor 110 is coupled to the first switching node SW1 and toa second switching node SW2. In particular, the first switching node SW1is configured to couple to a first terminal of the inductor 110 and thesecond switching node SW2 is configured to couple to a second terminalof the inductor 110. The third transistor 106 is coupled to the secondswitching node SW2 and to the output terminal 109, while the fourthtransistor 108 is coupled to the second switching node SW2 and to theground terminal 107. The output capacitor 114 is coupled between theoutput terminal 109 and the ground terminal 107.

In the example of FIG. 1, the controller 103 includes at least a statemachine 120 that is configured to control gate drivers 116, 118 tocontrol the transistors 102, 104, 106, 108 (e.g., to be conducting ornot conducting) of the voltage converter 101 to provide a desired VOUTfor a given VIN. The gate drivers 116, 118 may include charge pumps,which are not shown for simplicity. The state machine 120 is coupled tothe gates, for example, of first and second transistors 102, 104 by wayof the gate driver 116, and is coupled to the gates, for example, ofthird and fourth transistors 106, 108 by way of the gate driver 118.Although the gate drivers 116, 118 are shown as two separate modules forschematic simplicity, in some examples the functionality of the gatedrivers 116, 118 is carried out by more (e.g., one gate driver pertransistor) or fewer (e.g., one gate driver for all four transistors)modules. State machine 120 may be implemented, in some exampleembodiments, as a separate processing unit from controller 103 or partof a larger processing device. State machine 120 (and controller 103)may be implemented, in some example embodiments, using a processor (suchas a microprocessor or a microcontroller) or anapplication-specific-integrated-circuit (ASIC). For simplicity, it isassumed that the state machine 120, through the gate drivers 116, 118,controls or causes the voltage converter 101 to operate in various modes(e.g., buck mode, boost mode, or buck-boost mode) as described infurther detail below.

In the example of FIG. 1, the controller 103 also includes a firstcomparator 122 having an inverting terminal coupled to the outputterminal 109 and a non-inverting terminal configured to receive areference or threshold voltage (VREF). The first comparator 122 thuscompares VOUT to VREF and asserts its output (COMP OUT) in response toVOUT being less than VREF. The output of the first comparator 122 is aninput to the state machine 120, the function of which is described infurther detail below. The controller 103 is thus configured to detectVOUT based on the output of the first comparator 122.

The controller 103 also includes a second comparator 126 having anon-inverting terminal coupled to switching node SW1 and an invertingterminal configured to receive an upper current threshold referencevoltage (I_PEAK TARGET). Switching node SW1 is a schematicrepresentation of a node having a voltage that is related to (e.g.,proportional to) the current through the inductor 110 (IL), such as avoltage across a current sense resistor (or across one of thetransistors that is conducting) in series with the inductor 110, whichis not shown in FIG. 1 for simplicity. I_PEAK TARGET is related to(e.g., proportional to) an upper current threshold (I_PEAK), which isdescribed in further detail below. The second comparator 126 thuscompares IL to I_PEAK (or voltages proportional to IL and I_PEAK) andasserts its output in response to IL being greater than I_PEAK. Theoutput of the second comparator 126 (I_PEAK) is an input to the statemachine 120, the function of which is described in further detail below.

The controller 103 also includes a third comparator 128 having aninverting terminal coupled to switching node SW2 and a non-invertingterminal configured to receive a lower current threshold referencevoltage (I_VALLEY TARGET). As above, the switching node SW2 is aschematic representation of a node having a voltage that is related to(e.g., proportional to) IL. I_VALLEY TARGET is related to (e.g.,proportional to) a lower current threshold (I_VALLEY), which isdescribed in further detail below. The third comparator 128 thuscompares IL to I_VALLEY (or voltages proportional to IL and I_VALLEY)and asserts its output (I_VALLEY) in response to IL being less thanI_VALLEY. The output of the third comparator 128 is an input to thestate machine 120, the function of which is described in further detailbelow. The controller 103 is thus configured to detect IL based on theoutputs of the comparators 126, 128.

The controller 103 also includes a timer 124 (e.g., a counter) that iscoupled to the state machine 120. The state machine 120 supplies aninput to the timer 124 (e.g., to start the timer 124 in response to acondition being met). The state machine 120 also receives an input fromthe timer 124 (e.g., indicating a certain amount of time has elapsed).In some examples, the timer 124 also receives VIN and VOUT as inputs,which are used to determine an amount of time that the timer 124 isconfigured to indicate. The function of the timer 124 and the statemachine 120 are described in further detail below.

FIGS. 2-5 show the voltage converter 101 in various phases of itsoperation, being controlled by the controller 103 including the statemachine 120 described above, in accordance with various examples. Asexplained further below, operating the voltage converter 101 in the fourphases shown in FIGS. 2-5 allows the voltage converter 101 to operate inbuck mode, boost mode, or buck-boost mode while reducing the regulationand RHP zero issues described above. Additionally, and as explainedfurther below, a conversion energy of the voltage converter 101 iscontrolled by the controller 103 regulating the values of I_PEAK and/orI_VALLEY and the length of the various phases, described below.Regardless of whether the voltage converter 101 functions as a buckconverter, a boost converter, or a buck-boost converter, the voltageconverter 101 is controlled by the state machine 120 to cycle throughthe various phases described below.

In particular, FIG. 2 shows the voltage converter 101 in a boost-onphase. In the boost-on phase, the first transistor 102 and the fourthtransistor 108 are conducting, while the second transistor 104 and thethird transistor 106 are not conducting. As a result, a current path isformed as shown by the arrow in FIG. 2, and the inductor 110 isenergized by VIN. During the boost-on phase, the output capacitor 114provides energy (stored prior to this phase) to the output signal (e.g.,VOUT).

FIG. 3 shows the voltage converter 101 in a boost-off, buck-on phase. Inthe boost-off, buck-on phase, the first transistor 102 and the thirdtransistor 106 are conducting, while the second transistor 104 and thefourth transistor 108 are not conducting. As a result, a current path isformed as shown by the arrow in FIG. 3, and the input terminal 105 iscoupled to the output terminal 109 by the inductor 110.

FIG. 4 shows the voltage converter 101 in a buck-off phase. In thebuck-off phase, the second transistor 104 and the third transistor 106are conducting, while the first transistor 102 and the fourth transistor108 are not conducting. As a result, a current path is formed as shownby the arrow in FIG. 4, and the inductor 110 is de-energized byproviding energy to the output signal (e.g., VOUT). During the buck-offphase, the input capacitor 112 is charged by the input signal (e.g.,VIN).

FIG. 5 shows the voltage converter 101 in a pause phase. In the pausephase, the second transistor 104 and the fourth transistor 108 areconducting, while the first transistor 102 and the third transistor 106are not conducting. As a result, a current path is formed as shown bythe arrow in FIG. 5. In the pause phase, energy is preserved in thevoltage converter 101 because the inductor 110 is shorted, which resultsin an approximately constant current through the loop shown in FIG. 5,which decreases slightly according to a time constant of the inductor110 in combination with resistive losses. As explained further below, insome examples the state machine 120 causes the voltage converter 101 toremain in the pause phase as long as VOUT is greater than a target VOUTthreshold voltage. The introduction of the pause phase in examples ofthis disclosure allows energy flow to VOUT to be decreased whileavoiding adding energy to the voltage converter 101 (e.g., by energizingthe inductor 110 by VIN, as shown in FIG. 2). As a result, a balance ismaintained between the energy delivered to the voltage converter 101 andthe energy consumed at the output (e.g., by supplying VOUT) that isindependent of the magnitude of VIN and VOUT. As a result, distortionsdue to regulation activity are reduced. As explained further below,regulation of the voltage converter 101 is achieved through regulatingthe duration of the pause phase, while the energy supplied by thevoltage converter 101 in a pulse (e.g., a single cycle through the abovephases of FIGS. 2-5) is determined by I_PEAK, I_VALLEY, and the lengthof the boost-off, buck-on phase shown in FIG. 3.

FIG. 6 shows a state diagram 600 that illustrates the operation of thestate machine 120 as the controller 103 for the voltage converter 101,described above. The state diagram 600 includes a state 602 thatcorresponds to the state machine 120 controlling the voltage converter101 in the boost-on phase, described above with respect to FIG. 2. Thestate diagram 600 also includes a state 604 that corresponds to thestate machine 120 controlling the voltage converter 101 in theboost-off, buck-on phase, described above with respect to FIG. 3. Thestate diagram 600 further includes a state 606 that corresponds to thestate machine 120 controlling the voltage converter 101 in the buck-offphase, described above with respect to FIG. 4. Finally, the statediagram 600 includes a state 608 that corresponds to the state machine120 controlling the voltage converter 101 in the pause phase, describedabove with respect to FIG. 5.

During the state 602, in which the state machine 120 controls thevoltage converter 101 in the boost-on phase, the inductor 110 isenergized as VIN is applied across the inductor 110, causing the currentthrough the inductor 110 (IL) to increase. As a result of the secondcomparator 126 detecting that IL is greater than I_PEAK, the secondcomparator 126 output being asserted causes the state machine 120 totransition to the state 604.

During the state 604, in which the state machine 120 controls thevoltage converter 101 in the boost-off, buck-on phase, the inductor iscoupled to both the input terminal 105 and the output terminal 109. Inexamples in which the voltage converter 101 operates in buck mode, inwhich VOUT is less than VIN, IL continues to increase while the statemachine 120 operates in state 604 due to the polarity of the voltageacross the inductor 110 remaining similar as in state 602. However, inexamples in which the voltage converter 101 operates in boost mode, inwhich VOUT is greater than VIN, IL begins to decrease while the statemachine 120 operates in state 604 due to the polarity of the voltageacross the inductor 110 reversing relative to state 602. Similarly, inexamples in which the voltage converter 101 operates in buck-boost mode,in which VOUT is approximately equal to VIN, IL also begins to decreasewhile the state machine 120 operates in state 604 due to real-worldimpacts of non-ideal circuit behavior, such as resistive losses in boththe inductor 110 and the transistors 102, 106.

In response to entering the state 604, the state machine 120 isconfigured to signal to the timer 124 (e.g., by asserting a signalprovided to the timer 124) to begin timing. In response to the voltageconverter 101 being operated in buck mode, the timer 124 is configuredwith a time threshold (e.g., T_max) that is decreased in proportion tothe difference between VIN and VOUT (e.g., T_max=t0−k*(VIN−VOUT)). Thishas the effect of remaining in state 604, in which energy is transferredto the output (e.g., VOUT) from the inductor 110, for less time as thedifference between VIN and VOUT increases. This also reduces theinductor 110 ripple current, which would otherwise increase with longervalues of T_max and a large voltage across the inductor 110 (e.g.,VIN−VOUT). In response to the voltage converter 101 being operated inboost mode or buck-boost mode, the timer 124 is configured with a timethreshold, T_max=t0. In these examples, t0 or T_max are values that arerelated to the switching frequency of the voltage converter 101. Thetimer 124 output being asserted indicates that the time (t) kept by thetimer 124 is greater than T_max. Regardless of the mode of operation ofthe voltage converter 101 (e.g., boost mode, buck mode, or buck-boostmode), the timer 124 output being asserted causes the state machine 120to transition to the state 606.

During the state 606, in which the state machine 120 controls thevoltage converter 101 in the buck-off phase, the inductor 110 isde-energized by providing energy to the output signal (e.g., VOUT),causing IL to decrease. As a result of the third comparator 128detecting that IL is less than I_VALLEY, the third comparator 128 outputbeing asserted causes the state machine 120 to transition to the state608.

During the state 608, in which the state machine 120 controls thevoltage converter 101 in the pause phase, the energy is preserved in thevoltage converter 101 by shorting the inductor 110. While IL decreasesslightly due to the time constant of the inductor 110 and resistivelosses across the short circuit path, IL remains relatively stableduring the pause phase. As a result of the third comparator 128detecting that IL is less than I_VALLEY, the third comparator 128 outputbeing asserted causes the state machine 120 to transition to the state608. The state machine 120 remains in the state 608 until VOUT is lessthan the reference or threshold voltage (VREF). Thus, the regulation ofVOUT is through regulating the duration that the state machine 120remains in state 608. As a result of the first comparator 122 detectingthat VOUT is less than VREF, the first comparator 122 output beingasserted causes the state machine 120 to transition back to the state602.

Additionally, referring back to the state 604, in boost mode andbuck-boost mode, IL decreases as explained above. As a result of thethird comparator 128 detecting that IL is less than I_VALLEY, the thirdcomparator 128 output being asserted causes the state machine 120 totransition to the state 608. In an example, IL being less than I_VALLEYduring the state 604 is an indication that the voltage converter 101 hasalready provided more energy to the output signal (e.g., VOUT) thandesired for a given set of operating parameters. As a result, instead oftransitioning first to the state 606, in which the voltage converter 101provides additional energy to the output signal (e.g., VOUT), the statemachine 120 transitions directly to the state 608, in which the voltageconverter 101 energy is preserved. Subsequently, the state machine 120transitions back to state 602 as described above, and energy is againprovided to the voltage converter 101 by the input signal (e.g., VIN).

FIG. 7 shows a waveform 700 of IL as a function of time for the voltageconverter 101 operating in buck mode (VOUT<VIN) in accordance withvarious examples. The waveform 700 begins with the state machine 120operating in state 602 (e.g., boost-on phase) in which IL increases dueto VIN being applied across the inductor 110. At time 702, IL reachesI_PEAK, which causes the state machine 120 to transition to state 604 asexplained above. In this buck mode example, IL continues to increase,although at a slower rate, due to the voltage across the inductor 110(VIN−VOUT). At time 704, the timer 124 reaches T_max as explained above,which causes the state machine 120 to transition to state 606. IL thusbegins to decrease as the inductor 110 is de-energized by providingenergy to the output signal (e.g., VOUT). At time 706, IL reachesI_VALLEY, which causes the state machine 120 to transition to state 608as explained above. IL decreases slightly as a result of the inductor110 time constant and resistive losses across the short circuit path,but energy is generally preserved in the voltage converter 101 from time706 to 708. At time 708, VOUT reaches VREF, which causes the statemachine 120 to transition back to state 602, and the described cycle isrepeated.

FIG. 8 shows a waveform 800 of IL as a function of time for the voltageconverter 101 operating in boost mode (VOUT>VIN) in accordance withvarious examples. The waveform 800 begins with the state machine 120operating in state 602 (e.g., boost-on phase) in which IL increases dueto VIN being applied across the inductor 110. At time 802, IL reachesI_PEAK, which causes the state machine 120 to transition to state 604 asexplained above. In this boost mode example, IL begins to decrease, dueto the voltage across the inductor 110 reversing polarity (e.g.,VOUT>VIN). At time 804, the timer 124 time reaches T_max as explainedabove, which causes the state machine 120 to transition to state 606. ILthus continues to decrease as the inductor 110 is de-energized byproviding energy to the output (e.g., VOUT). At time 806, IL reachesI_VALLEY, which causes the state machine 120 to transition to state 608as explained above. IL decreases slightly as a result of a voltageacross the inductor 110, but energy is generally preserved in thevoltage converter 101 (via inductor 110) from time 806 to 808. At time808, VOUT reaches VREF, which causes the state machine 120 to transitionback to state 602, and the described cycle is repeated. Although notshown in the example of FIG. 8, in some examples, IL decreases morerapidly following time 802, and thus reaches I_VALLEY prior to the timer124 expiring. In such an example, the state machine 120 transitionsdirectly from state 604 to state 608, as explained above.

FIG. 9 shows a waveform 900 of IL as a function of time for the voltageconverter 101 operating in buck-boost mode (VOUT=VIN) in accordance withvarious examples. The waveform 900 begins with the state machine 120operating in state 602 (e.g., boost-on phase) in which IL increases dueto VIN being applied across the inductor 110. At time 902, IL reachesI_PEAK, which causes the state machine 120 to transition to state 604 asexplained above. In this buck-boost mode example, IL begins to decreaserelatively slowly, due to a relatively small voltage across the inductor110, because VOUT is approximately equal to VIN. At time 904, the timer124 time reaches T_max as explained above, which causes the statemachine 120 to transition to state 606. IL thus begins to decrease morerapidly as the inductor 110 is de-energized by providing energy to theoutput signal (e.g., VOUT). At time 906, IL reaches I_VALLEY, whichcauses the state machine 120 to transition to state 608 as explainedabove. IL decreases slightly as a result of the inductor 110 timeconstant and resistive losses across the short circuit path, but energyis generally preserved in the voltage converter 101 from time 906 to908. At time 908, VOUT reaches VREF, which causes the state machine 120to transition back to state 602, and the described cycle is repeated.Although not shown in the example of FIG. 9, in some examples ILdecreases more rapidly following time 902, and thus reaches I_VALLEYprior to the timer 124 expiring. In such an example, the state machine120 transitions directly from state 604 to state 608, as explainedabove.

In addition to the controller 103 described above, which uses the statemachine 120 to control the transistors 102, 104, 106, 108 of the voltageconverter 101, other examples of this disclosure relate to a controllerconfigured to regulate the conversion energy of a voltage converter.Such controllers often rely on an analog-to-digital converter (ADC) todigitize the analog value of VOUT, which is then processed by a digitalsignal processor (DSP) to appropriately control the conversion energy(e.g., the magnitude of I_PEAK and I_VALLEY) of the voltage converter.The use of such ADCs and DSPs is both complex and consumes a relativelylarge amount of power.

FIG. 10 shows a block diagram of a system 1000 in accordance withvarious examples. The system 1000 includes a voltage converter 1002 anda controller 1003 coupled to the voltage converter 1002. In an example,the voltage converter 1002 is a DC-DC converter such as a buck-boostconverter that converts an input voltage (VIN) at an input terminal toan output voltage (VOUT) at an output terminal. In at least someexamples, the voltage converter 1002 is configured to operate inmultiple modes (e.g., buck mode, boost mode, or buck-boost mode). In anexample, the voltage converter 1002 is structurally similar to thevoltage converter 101 descried above.

In the example of FIG. 10, the controller 1003 is configured to regulatethe conversion energy of the voltage converter 1002. Conversion energygenerally refers to the current level in the voltage converter 1002. Forexample, a greater current level results in more energy beingtransferred from the input signal (e.g., VIN) to the output signal(e.g., VOUT). In the particular example where the voltage converter 1002functions according to the above-described examples, the current levelof the voltage converter 1002 is affected by controlling the values ofI_PEAK and/or I_VALLEY. For example, increasing the values of I_PEAKand/or I_VALLEY increases the current level of the voltage converter1002, while decreasing the values of I_PEAK and/or I_VALLEY decreasesthe current level of the voltage converter 1002, as explained above.

In the example of FIG. 10, the controller 1003 includes a comparator1004 having an inverting terminal coupled to an output terminal of thevoltage converter 1002 (e.g., configured to receive VOUT) and anon-inverting terminal configured to receive a reference or thresholdvoltage (VREF). The comparator 1004 thus compares VOUT to VREF andasserts its output in response to VOUT being less than VREF. Referringto FIG. 6, above, VOUT being less than VREF satisfies the condition totransition from the state 608 to the state 602, which corresponds to atransition from the pause phase to the boost on phase. For the exampleof FIG. 10, this is referred to as the start of conversion, because theprevious conversion cycle ends with the conclusion of the pause phase.In an example, the comparator 1004 and the first comparator 122 areimplemented in a single component, the output of which is used by boththe state machine 120 and a timer 1006 of the controller 1003, asexplained further below.

The controller 1003 also includes the timer 1006 (e.g., a counter)having a start input (A) and a stop input (B1). In some examples, thetimer 1006 also has a disable input (B2) that, in response to beingasserted, causes the timer 1006 to turn off. The timer 1006 isconfigured to start timing in response to the start input being assertedand to stop timing in response to the stop input being asserted. Inresponse to the timer 1006 stopping, the timer 1006 is configured tolatch a time value (e.g., a digital counter value) as its output. Thestart input of the timer 1006 is coupled to the voltage converter 1002,which is asserted in response to the conclusion of the energy transferportion of a conversion cycle of the voltage converter 1002. In theexample of FIG. 6, the conclusion of the energy transfer portion of aconversion cycle occurs in response to the state machine 120transitioning to state 608 (e.g., from either state 604 or state 606).The stop input of the timer 1006 is coupled to the output of thecomparator 1004. As a result, the output of the timer 1006 correspondsto the duration of an energy preservation phase, such as the pause phasedescribed above, of the voltage converter 1002.

The controller 1003 also includes a time comparator 1008 coupled to thetimer 1006 and configured to receive the output of the timer 1006 as aninput. The time comparator 1008 is configured to receive a referencetime value (e.g., a digital value to which the output of the timer 1006is compared) as a second input. The time comparator 1008 includes aplurality of outputs. At a given time, one of the outputs of the timecomparator 1008 is asserted based on a relationship between the timer1006 output and reference time values input to the time comparator 1008.

For example, a first output of the time comparator 1008 is configured tobe asserted in response to the timer 1006 output being within a firstdeviation from the reference time value (e.g., TARGET+1−t(0)).Similarly, a second output of the time comparator 1008 is configured tobe asserted in response to the timer 1006 output being more than t(0)but less than a second deviation less than the reference time value(e.g., TARGET−t(1)). Further, a third output of the time comparator 1008is configured to be asserted in response to the timer 1006 output beingmore than t(0) but less than a third deviation greater than thereference time value (e.g., TARGET+t(2)). In some examples, the timecomparator 1008 includes additional outputs such as a fourth output thatis configured to be asserted in response to the timer 1006 output beingmore than t(1) less than the reference time value (e.g., TARGET−t(3)),and a fifth output that is configured to be asserted in response to thetimer 1006 output being more than t(2) greater than the reference timevalue (e.g., TARGET+t(4)).

In this example, the time comparator 1008 effectively bins thedifference between the timer 1006 output, which corresponds to theduration of an energy preservation phase, such as the pause phasedescribed above, of the voltage converter 1002, and the reference timevalue, which may be determined based on the preservation of energy inthe voltage converter 1002 during the pause phase. As a result, theenergy delivery of the voltage converter 1002 occurs during the periodof time between conversion start and conversion finished. In someexamples, in order to reduce the current level, and thus the lossesduring the pause phase, and also to provide sufficient control headroomfor a variation of the pause phase duration, the reference time value isa fraction of the period of time between conversion start and conversionfinished. The combination of the timer 1006 that measures the durationof the pause phase (e.g., the time taken for VOUT to fall below VREF)and the time comparator 1008 to compare the actual duration (e.g., thetimer 1006 output) with the reference time value or duration returns adegree of error. As a result, information regarding the voltage error ofVOUT is transferred into the time domain.

The controller 1003 further includes an accumulator 1010. The outputs ofthe time comparator 1008 are provided as inputs to the accumulator 1010.Thus, the accumulator 1010 is configured to be controlled by the binnedor classified error information from the time comparator 1008. Theoutput of the accumulator 1010 is a value that controls a level ofconversion energy of the voltage converter 1002. For example, anincrease in the accumulator 1010 output value results in an increase tothe values of I_PEAK and/or I_VALLEY. Continuing this example, adecrease in the accumulator 1010 output value results in a decrease tothe values of I_PEAK and/or I_VALLEY, as explained above.

In the example of FIG. 10, the accumulator 1010 is configured tomaintain its output value in response to the first output of the timecomparator 1008 being asserted. As explained above, the first output ofthe time comparator 1008 is asserted in response to the timer 1006output duration being within the first deviation t(0) of the referencetime duration. This indicates that the conversion energy for the voltageconverter 1002 (e.g., the I_PEAK and/or I_VALLEY values) is appropriatefor a particular load, and the accumulator 1010 output value, and thusthe conversion energy for the voltage converter 1002, is maintained.

The accumulator 1010 is configured to increase its output value inresponse to the second output of the time comparator 1008 beingasserted. As explained above, the second output of the time comparator1008 is asserted in response to the timer 1006 output duration beingmore than t(0) but less than TARGET−t(1) less than the reference timeduration. This indicates that the conversion energy for the voltageconverter 1002 is too low for a particular load (e.g., which results inthe pause phase being shorter than expected), and the accumulator 1010output value, and thus the conversion energy for the voltage converter1002, is increased.

The accumulator 1010 is configured to decrease its output value inresponse to the third output of the time comparator 1008 being asserted.As explained above, the third output of the time comparator 1008 isasserted in response to the timer 1006 output duration being more thant(0) but less than TARGET+t(2) greater than the reference time duration.This indicates that the conversion energy for the voltage converter 1002is too high for a particular load (e.g., which results in the pausephase being longer than expected), and the accumulator 1010 outputvalue, and thus the conversion energy for the voltage converter 1002, isdecreased.

In certain examples, the time comparator 1008 includes additionaloutputs, such as the fourth and fifth outputs explained above and shownin FIG. 10. In these examples, the amount by which the accumulator 1010increases or decreases its output value may vary depending on which oneof the time comparator 1008 outputs is asserted. For example, if thesecond output is asserted, the accumulator 1010 is configured toincrease its output value by a first amount (e.g., a value of 1 in theexample of FIG. 10). If the third output is asserted, the accumulator1010 is configured to decrease its output value by a second amount(e.g., also a value of 1 in the example of FIG. 10). However, if thefourth output is asserted, this indicates a larger error value becausethe pause phase duration was further below the reference time value thanexpected (e.g., less than TARGET−t(1)). Similarly, if the fifth outputis asserted, this indicates a larger error value because the pause phaseduration was even greater relative to the reference time value thanexpected (e.g., greater than TARGET+t(2)). In some examples, theaccumulator 1010 is configured to increase or decrease its output by agreater amount (e.g., +X or −Y) in response to the fourth or fifthoutputs, respectively, being asserted. This allows the accumulator 1010to more rapidly increase or decrease the conversion energy of thevoltage converter 1002 as needed.

In some examples, the accumulator 1010 is configured to increase thevalue of X in response to the fourth output of the time comparator 1008being asserted for multiple cycles in a row. Additionally, to furtherreduce power consumption of the controller 1003 in response to thevoltage converter 1002 supplying a light load, the fifth output of thetime comparator 1008 is also coupled to the disable input of the timer1006. Thus, in response to the pause phase being longer thanTARGET+t(2), the timer 1006 is also disabled to reduce powerconsumption.

In the example of FIG. 10, the controller 1003 regulates the voltageconverter 1002 operation using a one-bit ADC in the form of thecomparator 1004 and subsequent circuitry that operates in the timedomain in response to the value output by the timer 1006. As a result,in some examples, the power consumption of the controller 1003 is lessthan that of a controller that uses higher precision ADCs to digitizethe analog voltage value VOUT and then process the digitized voltagevalue in order to control the operation of a voltage converter.

FIG. 11 shows a state diagram 1100 of operation of the controller 1003shown in FIG. 10 in accordance with various examples. The state diagram1100 includes a state 1102 in which the voltage converter 1002 begins aconversion cycle (e.g., as a result of the output of the comparator 1004being asserted). The timer 1006 is also stopped in response to theconversion cycle beginning in the state 1102. The state diagram 1100then transitions to and remains in state 1104 until the energy transferportion of the conversion cycle is finished, for example as indicated bythe voltage converter 1002.

In response to the voltage converter 1002 asserting that the energytransfer portion of the conversion cycle is finished, the state diagram1100 transitions to the state 1106, in which the timer 1006 is clearedand started. As explained above, the start input of the timer 1006 iscoupled to an output of the voltage converter 1002 that is asserted inresponse to the energy transfer portion of the conversion cycle beingfinished.

After the timer 1006 is started in the state 1106, the state diagram1100 proceeds to state 1108 where it is determined whether VOUT is lessthan VREF (e.g., by the comparator 1004). While VOUT is greater thanVREF, the state diagram 1100 proceeds to block 1110 where it isdetermined whether the timer 1006 value is greater than a thirddeviation (e.g., t(4)) greater than a reference time value (e.g.,TARGET). While the timer 1006 value is less than TARGET+t(4), the statediagram 1100 returns to the state 1108. However, if the timer 1006 valueis greater than TARGET+t(4), the state diagram 1100 continues to thestate 1112 in which the timer 1006 is stopped or disabled (e.g., to savepower as explained above), at which point the state diagram 1100 alsoreturns to the state 1108 to determine when VOUT is less than VREF.

From the state 1108, in response to VOUT being less than VREF (e.g.,indicated by the comparator 1004 output), the state diagram 1100continues to the state 1114 where the timer 1006 is stopped. The statediagram 1100 then continues to the state 1116 in which the time valueoutput by the timer 1006 is compared with various thresholds. Asexplained above, if the timer 1006 output is within a first deviationfrom the reference time value (e.g., TARGET+/−t(0)), conversion energy(e.g., the values of I_PEAK and/or I_VALLEY) is maintained and thus thestate diagram 1100 returns to the state 1102 and a new conversion cyclebegins.

Referring back to the state 1116, if the timer 1006 output is more thant(0) but less than a second deviation less than the reference time value(e.g., TARGET−t(1)), the state diagram 1100 proceeds to the state 1120in which the conversion energy of the voltage converter 1002 isincreased by a first amount (e.g., 1). The state diagram 1100 thenreturns to the state 1102 and a new conversion cycle begins, in whichthe values of I_PEAK and/or I_VALLEY are increased relative to theirprevious values.

Referring back to the state 1116, if the timer 1006 output is more thant(0) but less than a third deviation greater than the reference timevalue (e.g., TARGET+t(2)), the state diagram 1100 proceeds to the state1122 in which the conversion energy of the voltage converter 1002 isdecreased by a second amount (e.g., 1). The state diagram 1100 thenreturns to the state 1102 and a new conversion cycle begins, in whichthe values of I_PEAK and/or I_VALLEY are decreased relative to theirprevious values.

Referring back to the state 1116, if the timer 1006 output is more thant(1) less than the reference time value (e.g., TARGET−t(3)), the statediagram 1100 proceeds to the state 1118 in which the conversion energyof the voltage converter 1002 is increased by a fourth amount (e.g., X).The state diagram 1100 then returns to the state 1102 and a newconversion cycle begins, in which the values of I_PEAK and/or I_VALLEYare further increased (e.g., X>1) relative to their previous values.

Again referring back to the state 1116, if the timer 1006 output is morethan t(2) greater than the reference time value (e.g., TARGET+t(4)), thestate diagram 1100 proceeds to the state 1124 in which the conversionenergy of the voltage converter 1002 is decreased by a fifth amount(e.g., Y). The state diagram 1100 then returns to the state 1102 and anew conversion cycle begins, in which the values of I_PEAK and/orI_VALLEY are further decreased (e.g., Y>1) relative to their previousvalues.

As explained above, the state diagram 1100 provides a method to regulatethe voltage converter 1002 operation using a one-bit ADC in the form ofthe comparator 1004 and subsequent circuitry that operates in the timedomain in response to the value output by the timer 1006. As a result,in some examples, the power consumption of the controller 1003 thatimplements the state diagram 1100 is less than that of a controller thatuses higher precision ADCs to digitize the analog voltage value VOUT andthen process the digitized voltage value in order to control theoperation of a voltage converter.

FIG. 12 shows waveforms 1200 of VOUT, inductor current (IL), andaccumulator 1010 output as a function of time for the voltage converter101, 1002 operating in the buck mode in accordance with variousexamples. In particular, in response to determining that the pause phaseis shorter than the reference time value (PAUSE<TARGET), the accumulator1010 output is increased from a value of 0×56 to a value of 0×57. As aresult, the conversion energy of the voltage converter 101, 1002 isincreased by increasing the value of I_PEAK. Subsequently, in responseto determining that the pause phase is within a first deviation of thereference time value (PAUSE=TARGET), the accumulator 1010 output ismaintained at a value of 0×57. Finally, in response to determining thatthe pause phase is longer than the reference time value (PAUSE>TARGET),the accumulator 1010 output is decreased from 0×57 back to 0×56. Theforegoing is one example and it is noted that such regulation continues,with varying changes to the accumulator 1010 output as described abovewith respect to FIGS. 10 and 11.

FIG. 13 shows waveforms 1300 of VOUT, IL, and accumulator 1010 output asa function of time for the voltage converter 101, 1002 operating in thebuck-boost mode in accordance with various examples. The waveforms 1300are generally similar to the waveforms 1200 described above. Forexample, in response to determining that the pause phase is shorter thanthe reference time value, the accumulator 1010 output is increased froma value of 0×1b to a value of 0×1c. As a result, the conversion energyof the voltage converter 101, 1002 is increased by increasing the valueof I_PEAK. Subsequently, in response to determining that the pause phaseis longer than the reference time value, the accumulator 1010 output isdecreased from 0×1c back to 0×1b. During the next cycle, in response todetermining that the pause phase is still longer than the reference timevalue (e.g., decreasing the accumulator 1010 output by 1 wasinsufficient to shorten the pause phase to the desired duration), theaccumulator 1010 output is decreased further to 0×1a. As a result, theconversion energy of the voltage converter 101, 1002 is decreased bydecreasing the value of I_PEAK correspondingly with the decrease of theaccumulator 1010 output. The foregoing is one example and it is notedthat such regulation continues, with varying changes to the accumulator1010 output as described above with respect to FIGS. 10 and 11.

FIG. 14 shows waveforms 1400 of VOUT, IL, and accumulator 1010 output asa function of time for the voltage converter 101, 1002 operating in theboost mode in accordance with various examples. The waveforms 1400 aregenerally similar to the waveforms 1200, 1300 described above. Forexample, in response to determining that the pause phase is longer thanthe reference time value, the accumulator 1010 output is decreased froma value of 0×25 to a value of 0×24. As a result, the conversion energyof the voltage converter 101, 1002 is decreased by decreasing the valueof I_PEAK. Subsequently, in response to determining that the pause phaseis shorter than the reference time value, the accumulator 1010 output isincreased from 0×24 back to 0×25. This behavior continues to regulatethe length of the pause phase. The foregoing is one example and it isnoted that such regulation continues, with varying changes to theaccumulator 1010 output as described above with respect to FIGS. 10 and11.

In the foregoing discussion, the terms “including” and “comprising” areused in an open-ended fashion, and thus should be interpreted to mean“including, but not limited to . . . .” The term “couple” is usedthroughout the specification. The term may cover connections,communications, or signal paths that enable a functional relationshipconsistent with the description of the present disclosure. For example,if device A generates a signal to control device B to perform an action,in a first example device A is coupled to device B, or in a secondexample device A is coupled to device B through intervening component Cif intervening component C does not substantially alter the functionalrelationship between device A and device B such that device B iscontrolled by device A via the control signal generated by device A. Adevice that is “configured to” perform a task or function may beconfigured (e.g., programmed and/or hardwired) at a time ofmanufacturing by a manufacturer to perform the function and/or may beconfigurable (or re-configurable) by a user after manufacturing toperform the function and/or other additional or alternative functions.The configuring may be through firmware and/or software programming ofthe device, through a construction and/or layout of hardware componentsand interconnections of the device, or a combination thereof.Furthermore, a circuit or device that is said to include certaincomponents may instead be configured to couple to those components toform the described circuitry or device. For example, a structuredescribed as including one or more semiconductor elements (such astransistors), one or more passive elements (such as resistors,capacitors, and/or inductors), and/or one or more sources (such asvoltage and/or current sources) may instead include only thesemiconductor elements within a single physical device (e.g., asemiconductor die and/or integrated circuit (IC) package) and may beconfigured to couple to at least some of the passive elements and/or thesources to form the described structure either at a time of manufactureor after a time of manufacture, for example, by an end-user and/or athird-party.

What is claimed is:
 1. A system having an input terminal and an outputterminal, the system comprising: a voltage converter, including: a firsttransistor coupled to the input terminal and to a first switching node;a second transistor coupled to the first switching node and to a groundnode; a third transistor coupled to a second switching node and to theoutput terminal; a fourth transistor coupled to the second switchingnode and to the ground node; and an inductor having a first terminalcoupled to the first switching node and a second terminal coupled to thesecond switching node; and a controller coupled to the voltageconverter, the controller comprising: a state machine; a plurality ofdrivers configured to control the transistors of the voltage converter;and wherein the state machine is adaptable to cause the second andfourth transistors to be conducting and the first and third transistorsto be non-conducting, in response to a current through the inductorbeing less than a current threshold.
 2. The system of claim 1, whereinthe controller further includes a comparator comprising: an invertinginput coupled to a reference voltage source that is proportional to thecurrent threshold; a non-inverting input coupled to an inductor node,wherein a voltage at the inductor node is proportional to the currentthrough the inductor; and an output.
 3. The system of claim 1, whereinthe state machine is adaptable to transition to cause the first andfourth transistors to be conducting and the second and third transistorsto be non-conducting.
 4. The system of claim 3, wherein the controllerfurther includes a comparator comprising: an inverting input coupled tothe output terminal; a non-inverting input coupled to a referencevoltage source; and an output.
 5. The system of claim 3, wherein: thecurrent threshold is a first current threshold; the state machine isadaptable to transition to a state in which the first and thirdtransistors are conducting and the second and fourth transistors arenon-conducting, in response to the current through the inductor beinggreater than a second current threshold; and the second currentthreshold is greater than the first current threshold.
 6. The system ofclaim 5, wherein the controller further includes a comparatorcomprising: an inverting input coupled to an inductor node, wherein avoltage at the inductor node is proportional to the current through theinductor; a non-inverting input coupled to a reference voltage source,wherein a voltage provided by the reference voltage source isproportional to the second current threshold; and an output.
 7. Thesystem of claim 5, wherein the state machine is adaptable to transitionto a state in which the second and third transistors are conducting andthe first and fourth transistors are non-conducting based on time in aprevious state.
 8. The system of claim 7, wherein the controller isfurther configured to control the voltage converter in a buck mode,wherein in the buck mode, the time is based on a voltage at the inputterminal and the voltage at the output terminal.
 9. The system of claim7, wherein the controller is further configured to control the voltageconverter in a boost mode or a buck-boost mode, wherein in the boostmode or the buck-boost mode, the time is a constant value.
 10. Acontroller for a voltage converter having a first switching node, asecond switching node and an inductor coupled between the first andsecond switching nodes, the controller comprising: a state machine; aplurality of drivers, each driver coupled to a transistor; and whereinthe state machine, in response to a current through the inductor beingless than a first current threshold, is adaptable to transition to astate in which: a first transistor between the first switching node andan input terminal is not conducting; a second transistor between thefirst switching node and a ground node is conducting; a third transistorbetween the second switching node and an output terminal is notconducting; and a fourth transistor between the second switching nodeand the ground node is conducting.
 11. The controller of claim 10,wherein the state machine is adaptable to transition to a state in whichthe first and fourth transistors are conducting and the second and thirdtransistors are not conducting.
 12. The controller of claim 11, whereinthe state machine is adaptable to transition to the state in which thefirst and fourth transistors are conducting and the second and thirdtransistors are not conducting in response to a voltage at the outputterminal being below a voltage threshold.
 13. The controller of claim11, wherein: the state machine, in response to the current through theinductor being greater than a second current threshold, is adaptable totransition to a state in which the first and third transistors areconducting and the second and fourth transistors are not conducting; andthe second current threshold is greater than the first currentthreshold.
 14. The controller of claim 13, wherein the state machine isadaptable to transition, based on time threshold in a previous state, toa state in which the second and third transistors are conducting and thefirst and fourth transistors are not conducting.
 15. The controller ofclaim 14, wherein the controller is further configured to control thevoltage converter in a buck mode, wherein in the buck mode, the timethreshold is based on a voltage at the input terminal and the voltage atthe output terminal.
 16. The controller of claim 14, wherein thecontroller is further configured to control the voltage converter in aboost mode or a buck-boost mode, wherein in the boost mode or thebuck-boost mode, the time threshold is a constant value.
 17. A methodfor controlling a voltage converter, the method comprising: detecting,by a controller, a current through an inductor coupled to a firstswitching node and to a second switching node of the voltage converter;in response to the current through the inductor being less than acurrent threshold: causing, by the controller, a first transistorbetween the first switching node and an input terminal not to conduct;causing, by the controller, a second transistor between the firstswitching node and a ground node to conduct; causing, by the controller,a third transistor between the second switching node and an outputterminal not to conduct; and causing, by the controller, a fourthtransistor between the second switching node and the ground node toconduct.
 18. The method of claim 17, further comprising: detecting, bythe controller, a voltage at the output terminal; in response to thevoltage at the output terminal being below a voltage threshold: causing,by the controller, the first transistor to conduct; causing, by thecontroller, the second transistor not to conduct; causing, by thecontroller, the third transistor not to conduct; and causing, by thecontroller, the fourth transistor to conduct.
 19. The method of claim18, wherein the current threshold is a first current threshold, themethod further comprising: in response to the current through theinductor being greater than a second current threshold: causing, by thecontroller, the first transistor to conduct; causing, by the controller,the second transistor not to conduct; causing, by the controller, thethird transistor to conduct; and causing, by the controller, the fourthtransistor not to conduct; wherein the second current threshold isgreater than the first current threshold.
 20. The method of claim 19,further comprising: determining an amount of time in which the first andthird transistors are conducting and the second and fourth transistorsare not conducting; in response to the amount of time being greater thana time threshold: causing, by the controller, the first transistor notto conduct; causing, by the controller, the second transistor toconduct; causing, by the controller, the third transistor to conduct;and causing, by the controller, the fourth transistor not to conduct.